Multipoint touch surface controller

ABSTRACT

A multipoint touch surface controller is disclosed herein. The controller includes an integrated circuit including output circuitry for driving a capacitive multi-touch sensor and input circuitry for reading the sensor. Also disclosed herein are various noise rejection and dynamic range enhancement techniques that permit the controller to be used with various sensors in various conditions without reconfiguring hardware.

BACKGROUND

There exist today many styles of input devices for performing operationsin a computer system. The operations generally correspond to moving acursor and/or making selections on a display screen. By way of example,the input devices may include buttons or keys, mice, trackballs, touchpads, joy sticks, touch screens and the like. Touch pads and touchscreens (collectively “touch surfaces” are becoming increasingly popularbecause of their ease and versatility of operation as well as to theirdeclining price. Touch surfaces allow a user to make selections and movea cursor by simply touching the surface, which may be a pad or thedisplay screen, with a finger, stylus, or the like. In general, thetouch surface recognizes the touch and position of the touch and thecomputer system interprets the touch and thereafter performs an actionbased on the touch.

Of particular interest are touch screens. Various types of touch screensare described in applicant's co-pending patent application Ser. No.10/840,862, entitled “Multipoint Touchscreen,” filed May 6, 2004, whichis hereby incorporated by reference in its entirety. As noted therein,touch screens typically include a touch panel, a controller and asoftware driver. The touch panel is generally a clear panel with a touchsensitive surface. The touch panel is positioned in front of a displayscreen so that the touch sensitive surface covers the viewable area ofthe display screen. The touch panel registers touch events and sendsthese signals to the controller. The controller processes these signalsand sends the data to the computer system. The software drivertranslates the touch events into computer events.

There are several types of touch screen technologies includingresistive, capacitive, infrared, surface acoustic wave, electromagnetic,near field imaging, etc. Each of these devices has advantages anddisadvantages that are taken into account when designing or configuringa touch screen. One problem found in these prior art technologies isthat they are only capable of reporting a single point even whenmultiple objects are placed on the sensing surface. That is, they lackthe ability to track multiple points of contact simultaneously. Inresistive and traditional capacitive technologies, an average of allsimultaneously occurring touch points are determined and a single pointwhich falls somewhere between the touch points is reported. In surfacewave and infrared technologies, it is impossible to discern the exactposition of multiple touch points that fall on the same horizontal orvertical lines due to masking. In either case, faulty results aregenerated.

These problems are particularly problematic in handheld devices, such astablet PCs, where one hand is used to hold the tablet and the other isused to generate touch events. For example, as shown in FIGS. 1A and 1B,holding a tablet 2 causes the thumb 3 to overlap the edge of the touchsensitive surface 4 of the touch screen 5. As shown in FIG. 1A, if thetouch technology uses averaging, the technique used by resistive andcapacitive panels, then a single point that falls somewhere between thethumb 3 of the left hand and the index finger 6 of the right hand wouldbe reported. As shown in FIG. 1B, if the technology uses projectionscanning, the technique used by infrared and surface acoustic wavepanels, it is hard to discern the exact vertical position of the indexfinger 6 due to the large vertical component of the thumb 3. The tablet2 can only resolve the patches shown in gray. In essence, the thumb 3masks out the vertical position of the index finger 6.

While virtually all commercially available touch screen based systemsavailable today provide single point detection only and have limitedresolution and speed, other products available today are able to detectmultiple touch points. Unfortunately, these products only work on opaquesurfaces because of the circuitry that must be placed behind theelectrode structure. Examples of such products include the Fingerworksseries of touch pad products. Historically, the number of pointsdetectable with such technology has been limited by the size of thedetection circuitry.

Therefore, what is needed in the art is a multi-touch capable touchscreen controller that facilitates the use of transparent touch sensorsand provides for a conveniently integrated package.

SUMMARY

A controller for multi-touch touch surfaces is disclosed herein. Oneaspect of the multi-touch touch surface controller relates to theintegration of drive electronics for stimulating the multi-touch sensorand sensing circuits for reading the multi-touch sensor into a singleintegrated package.

Another aspect of the controller relates to a technique for suppressingnoise in the sensor by providing a plurality of stimulus waveforms tothe sensor wherein the waveforms have different frequencies. Thispermits at least one noise-free detection cycle in cases where noiseappears at a particular frequency.

Another aspect of the controller relates to a charge amplifier thatincludes programmable components, namely, programmable resistors andcapacitors to allow the circuit to be easily reconfigured to provideoptimum sensing configurations for a variety of sensor conditions.

Another aspect of the controller relates to an offset compensationcircuit that expands the dynamic range of the controller by eliminatinga static portion of the multi-touch surface sensor output allowing thefull dynamic range of the sensing circuitry to be allocated to thechanging portions of the output signal.

Another aspect of the controller relates to a demodulation circuit thatenhances the noise immunity of the sensor arrangement by application ofparticular demodulation waveforms known to have particular frequencycharacteristics.

Another aspect of the controller relates to the application of variousalgorithms to the sensor outputs obtained from the multiple stimulusfrequencies described above to further increase noise immunity of thesystem.

These and other aspects will be more readily understood by reference tothe following detailed description and figures.

BRIEF DESCRIPTION OF THE DRAWINGS

FIGS. 1A and 1B illustrates certain problems with prior art touch screentechnologies.

FIG. 2 illustrates a perspective view of a computing deviceincorporating a multi-touch touch screen and multi-touch touch screencontroller according to certain teachings of the present disclosure.

FIG. 3 is a block diagram of a computing device incorporating amulti-touch touch screen and multi-touch touch screen controlleraccording to certain teachings of the present disclosure.

FIGS. 4A and 4B illustrate two possible arrangement of drive and senseelectrodes in a multi-touch touch surface.

FIG. 5 is a layer diagram illustrating communication between themulti-touch surface and the host computer device by way of a multi-touchcontroller incorporating various teachings of the present disclosure.

FIG. 6 is an equivalent circuit showing the output circuitry of thecontroller, a cell of the multi-touch sensor, and the input circuitry ofa multi-touch controller incorporating various teachings of the presentdisclosure.

FIG. 7 is a circuit schematic of a charge amplifier incorporated incertain embodiments of a multi-touch controller incorporating variousteachings of the present disclosure.

FIG. 8 is a block diagram of the multi-touch surface and multi-touchcontroller system in accordance with various teachings of the presentdisclosure.

FIG. 9 illustrates the sequence in which drive waveforms of varyingfrequencies are applied to the multi-touch sensor in accordance withcertain teachings of the present disclosure.

FIG. 10 is a block diagram illustrating the input circuitry of amulti-touch controller incorporating certain teachings of the presentdisclosure.

FIGS. 11A and 11B illustrate various demodulation waveforms togetherwith frequency spectra of their passbands.

FIG. 12 illustrates a sequence of stimulus waveforms, together with aparticular demodulation waveform, and the resulting output.

FIG. 13 illustrates the noise rejection technique employed by themajority rules algorithm.

FIG. 14 is a flowchart depicting exemplary steps in the operation of thecontroller in accordance with various teachings of the presentdisclosure.

DETAILED DESCRIPTION

A multipoint touch screen controller (MTC) is described herein. Thefollowing embodiments of the invention, described in terms of devicesand applications compatible with computer systems manufactured by AppleComputer, Inc. of Cupertino, Calif., are illustrative only and shouldnot be considered limiting in any respect.

FIG. 2 is a perspective view of a touch screen display arrangement 30.Display arrangement 30 includes a display 34 and a transparent touchscreen 36 positioned in front of display 34. Display 34 may beconfigured to display a graphical user interface (GUI) including perhapsa pointer or cursor as well as other information to the user.Transparent touch screen 36 is an input device that is sensitive to auser's touch, allowing a user to interact with the graphical userinterface on display 34. In general, touch screen 36 recognizes touchevents on surface 38 of touch screen 36 and thereafter outputs thisinformation to a host device. The host device may, for example,correspond to a computer such as a desktop, laptop, handheld or tabletcomputer. The host device interprets the touch event and thereafterperforms an action based on the touch event.

In contrast to prior art touch screens, touch screen 36 shown herein isconfigured to recognize multiple touch events that occur simultaneouslyat different locations on touch sensitive surface 38. That is, touchscreen 36 allows for multiple contact points T1-T4 to be trackedsimultaneously. Touch screen 36 generates separate tracking signalsS1-S4 for each touch point T1-T4 that occurs on the surface of touchscreen 36 at the same time. In one embodiment, the number ofrecognizable touches may be about 15 which allows for a user's 10fingers and two palms to be tracked along with 3 other contacts. Themultiple touch events can be used separately or together to performsingular or multiple actions in the host device. Numerous examples ofmultiple touch events used to control a host device are disclosed inU.S. Pat. Nos. 6,323,846; 6,888,536; 6,677,932; 6,570,557, andco-pending U.S. patent application Ser. Nos. 11/015,434; 10/903,964;11/048,264; 11/038,590; 11/228,758; 11/228,700; 11/228,737; 11/367,749,each of which is hereby incorporated by reference in its entirely.

FIG. 3 is a block diagram of a computer system 50, employing amulti-touch touch screen. Computer system 50 may be, for example, apersonal computer system such as a desktop, laptop, tablet, or handheldcomputer. The computer system could also be a public computer systemsuch as an information kiosk, automated teller machine (ATM), point ofsale machine (POS), industrial machine, gaming machine, arcade machine,vending machine, airline e-ticket terminal, restaurant reservationterminal, customer service station, library terminal, learning device,etc.

Computer system 50 includes a processor 56 configured to executeinstructions and to carry out operations associated with the computersystem 50. Computer code and data required by processor 56 are generallystored in storage block 58, which is operatively coupled to processor56. Storage block 58 may include read-only memory (ROM) 60, randomaccess memory (RAM) 62, hard disk drive 64, and/or removable storagemedia such as CD-ROM, PC-card, floppy disks, and magnetic tapes. Any ofthese storage devices may also be accessed over a network. Computersystem 50 also includes a display device 68 that is operatively coupledto the processor 56. Display device 68 may be any of a variety ofdisplay types including liquid crystal displays (e.g., active matrix,passive matrix, etc.), cathode ray tubes (CRT), plasma displays, etc.

Computer system 50 also includes touch screen 70, which is operativelycoupled to the processor 56 by I/O controller 66 and touch screencontroller 76. (The I/O controller 66 may be integrated with theprocessor 56, or it may be a separate component.) In any case, touchscreen 70 is a transparent panel that is positioned in front of thedisplay device 68, and may be integrated with the display device 68 orit may be a separate component. Touch screen 70 is configured to receiveinput from a user's touch and to send this information to the processor56. In most cases, touch screen 70 recognizes touches and the positionand magnitude of touches on its surface.

Better understanding of the interface between the touch sensor and thehost computer system may be had with reference to FIG. 5, which is alayer diagram of the system illustrated in FIG. 3. The touch sensor 301resides at the lowermost layer. In a preferred embodiment, the sensorinterfaces with an ASIC (application specific integrated circuit) 305that stimulates the sensor and reads the raw sensor output as describedin more detail below. ASIC 305 interfaces via signaling 306 with a DSP(digital signal processor) and/or microcontroller 307, which generatesthe capacitance images. Together ASIC 305 and DSP/microcontroller 307form the multipoint touch screen controller.

DSP/Microcontroller 307 includes an interface 308 for accepting thesignaling 306 from ASIC 305, and these signals are then passed to a datacapture and error rejection layer 309. Data from this layer may beaccessed both for calibration, baseline and standby processing by module310, as well as feature (i.e., touch point) extraction and compressionmodule 311. Once the features are extracted they are passed ashigh-level information to the host computer 302 via interface 303.Interface 303 may be, for example, a USB (universal serial bus)interface. Alternatively, other forms of interface, such as IEEE 1394(“Firewire”, RS-232 serial interface, SCSI (small computer systemsinterface), etc. could be used.

The exact physical construction of the sensing device is not necessaryfor a complete understanding touch screen controller disclosed herein.Nonetheless, details of the construction may be understood by referenceto the patents and patent applications incorporated by reference above.For purposes of the present description, the sensor may be assumed to bea mutual capacitance device constructed as described below withreference to FIGS. 4A and 4B.

The sensor panel is comprised of a two-layered electrode structure, withdriving lines on one layer and sensing lines on the other. In eithercase, the layers are separated by a dielectric material. In theCartesian arrangement of FIG. 4A, one layer is comprised of Nhorizontal, preferably equally spaced row electrodes 81, while the otherlayer is comprised of M vertical, preferably equally spaced columnelectrodes 82. In a polar arrangement, illustrated in FIG. 4B, thesensing lines may be concentric circles and the driving lines may beradially extending lines (or vice versa). As will be appreciated bythose skilled in the art, other configurations based on an infinitevariety of coordinate systems are also possible.

Each intersection 83 represents a pixel and has a characteristic mutualcapacitance, C_(SIG). A grounded object (such as a finger) thatapproaches a pixel 83 from a finite distance shunts the electric fieldbetween the row and column intersection, causing a decrease in themutual capacitance C_(SIG) at that location. In the case of a typicalsensor panel, the typical signal capacitance C_(SIG) is about 0.75 pFand the change induced by a finger touching a pixel, is about 0.25 pF.

The electrode material may vary depending on the application. In touchscreen applications, the electrode material may be ITO (Indium TinOxide) on a glass substrate. In a touch tablet, which need not betransparent, copper on an FR4 substrate may be used. The number ofsensing points 83 may also be widely varied. In touch screenapplications, the number of sensing points 83 generally depends on thedesired sensitivity as well as the desired transparency of the touchscreen 70. More nodes or sensing points generally increases sensitivity,but reduces transparency (and vice versa).

During operation, each row (or column) is sequentially charged bydriving it with a predetermined voltage waveform 85 (discussed ingreater detail below). The charge capacitively couples to the columns(or rows) at the intersection. The capacitance of each intersection 83is measured to determine the positions of multiple objects when theytouch the touch surface. Sensing circuitry monitors the chargetransferred and time required to detect changes in capacitance thatoccur at each node. The positions where changes occur and the magnitudeof those changes are used to identify and quantify the multiple touchevents. Driving each row and column and sensing the charge transfer isthe function of a multipoint touch screen controller.

FIG. 6 is a simplified diagram of the equivalent mutual capacitancecircuit 220 for each coupling node. Mutual capacitance circuit 220includes a driving line 222 and a sensing line 224 that are spatiallyseparated thereby forming a capacitive coupling node 226. When no objectis present, the capacitive coupling at node 226 stays fairly constant.When an object, such as a finger, is placed proximate the node 226, thecapacitive coupling through node 226 changes. The object effectivelyshunts the electric field so that the charge transferred across node 226is less.

With reference to FIGS. 5 and 8, ASIC 305 generates all the drivewaveforms necessary to scan the sensor panel. Specifically, themicroprocessor sends a clock signal 321 to set the timing of the ASIC,which in turn generates the appropriate timing waveforms 322 to createthe row stimuli to the sensor 301. Decoder 311 decodes the timingsignals to drive each row of sensor 301 in sequence. Level shifter 310converts timing signals 322 from the signaling level (e.g., 3.3V) to thelevel used to drive the sensor (e.g., 18V).

Each row of the sensor panel is driven determined by microprocessor 307.For noise rejection purposes it is desirable to drive the panel atmultiple different frequencies. Noise that exists at a particular drivefrequency may not, and likely will not exist at the other frequencies.In a preferred embodiment, each sensor panel row is stimulated withthree bursts of twelve square wave cycles (50% duty-cycle, 18Vamplitude), while the remaining rows are kept at ground. For betternoise rejection, described in greater detail below the frequency of eachburst is different, exemplary burst frequencies are 140 kHz, 200 kHz,and 260 Khz.

During each burst of pulses ASIC 305 takes a measurement of the columnelectrodes. This process is repeated for all remaining rows in thesensor panel. The results are three images, each image taken at adifferent stimulus frequency.

Additionally, it is preferable to minimize the amount of stimulusfrequency change required for each subsequent burst. Therefore afrequency hopping pattern that minimizes the changes is desirable. FIG.29 shows one possible frequency hopping pattern. In this arrangement, afirst row is driven with a 140 kHz burst, then a 200 kHz, and finally a260 kHz burst. Then a next row is driven with three bursts at 260 kHz,200 kHz, and 140 kHz, respectively. This particular frequency patternwas chosen to keep changes between frequencies small and allow thefrequency transitions have to be smooth and glitch free. However, otherfrequency hopping arrangements are also possible, including scanningmore than three frequencies, scanning the frequencies in a quasi-randomsequence rather than the ordered pattern described, and adaptivefrequency hopping, in which the scan frequencies are selected based onthe noise environment.

Turning back to FIG. 6, sensing line 224 is electrically coupled to acapacitive sensing circuit 230. Capacitive sensing circuit 230 detectsand quantifies the current change and the position of the node 226 wherethe current change occurred and reports this information to a hostcomputer. The signal of interest is the capacitance C_(SIG), whichcouples charge from RC network A to RC network B. The output from RCnetwork B connects directly to the analog input terminals of ASIC 305.ASIC 305 also uses the clock signal 321 (FIG. 8) from microprocessor 307(FIG. 8) to time the detection and quantification of the capacitancesignals.

FIG. 10 is a block diagram illustrating the input stage of ASIC 305. Theinput signal is first received by a charge amplifier 401. The chargeamplifier performs the following tasks: (1) charge to voltageconversion, (2) charge amplification, (3) rejection or stray capacitancepresent at the column electrode, and (4) anti aliasing, and (5) gainequalization at different frequencies. FIG. 7 is a diagram of onepossible charge amplifier 401.

Charge to voltage conversion is performed by a capacitor C_(FB) in thefeedback path of an operational amplifier 450. In one embodiment, thefeedback capacitor can be programmed with values ranging from 2 to 32pF, which allows the output voltage level to be adjusted to obtain thebest dynamic range for a range of C_(SIG) values. The feedback resistorR_(FB) is also preferably programmable to control the amplifier gain.

Because C_(SIG) will vary across a touch surface because of a variety ofmanufacturing tolerance related factors, it is useful to adjust thecharge amplifier feedback capacitance C_(FB) on a per-pixel basis. Thisallows gain compensation to be performed to optimize the performance ofeach pixel. In one embodiment, quasi-per pixel adjustment is performedas follows: The feedback capacitor C_(FB) has its value set by aregister known as CFB_REG. The value of CFB_REG is set according to thefollowing equation:CFB _(—) REG[Y]=CFB _(—) UNIV+CFB[Y]where Y is an individual pixel within a row, CFB_UNIV is adjusted on arow by row basis, and CFB[Y] is a lookup table loaded at system startup.In alternative arrangements, CFB_UNIV may be constant for all rows, orthe CFB[Y] lookup table may be switched out on a row by row basis. Also,although discussed in terms of rows and columns, the adjustmentarrangement is equally applicable to non-Cartesian coordinate systems.

Obviously it is desirable to measure C_(SIG) while rejecting as much aspossible the effects of any parasitic resistance and capacitance in thephysical sensor. Rejection of parasitic resistance and capacitance inthe sensor may be accomplished by holding the non-inverting input 451 ofamplifier 45D at a constant value, e.g., ground. The inverting input 452is coupled to the node being measured. As will be appreciated by thoseskilled in the art, inverting input 452 (connected to the columnelectrode being measured) is thus held at virtual ground. Therefore anyparasitic capacitance present at the column electrode, e.g., PCB straycapacitance or dynamic stray capacitance caused by the user touching thecolumn electrode, is rejected because the net charge of the straycapacitor does not change (i.e., the voltage across the straycapacitance is held at virtual ground). Therefore the charge amplifieroutput voltage 453 is only a function of the stimulus voltage, C_(SIG),and C_(FB). Because the stimulus voltage and C_(FB) are determined bythe controller, C_(SIG) may be readily inferred.

A series resistor 454 between the ASIC input pin 455 and the invertinginput 452 of the charge amplifier forms an anti-aliasing filter incombination with the feedback network of R_(FB) and C_(FB).

The high pass roll off of the charge amplifier is set by the parallelcombination of the feedback resistor R_(FB) and the feedback capacitorC_(FB).

Again with reference to FIG. 10, the output of charge amplifier 401passes to demodulator 403. Demodulator 403 is a 5-bit quantizedcontinuous time analog (four-quadrant) multiplier. The purpose ofdemodulator 403 is to reject out of band noise sources (from cellphones, microwave ovens, etc.) that are present on the signal enteringASIC 305. The output 402 of the charge amplifier (V_(SIG)) is mixed witha 5-bit quantized waveform that is stored in a lookup table 404. Theshape, amplitude, and frequency of the demodulation waveform isdetermined by programming suitable coefficients into lookup table 404.The demodulation waveform determines pass band, stop band, stop bandripple and other characteristics of the mixer. In a preferredembodiment, Gaussian shaped sine wave is used as the demodulationwaveform. A Gaussian sine wave provides a sharp pass band with reducedstop band ripple.

Another aspect of demodulator 403 relates to demodulator phase delayadjustment. As can be seen with reference to FIG. 10, the touch surfaceelectrodes can be represented by a RC networks (RC Network A and RCNetwork B) that have a mutual capacitance (C_(SIG)) at the point theyintersect. Each RC network constitutes a low pass filter, while C_(SIG)introduces a high pass filter response. Therefore the touch panel lookslike a bandpass filter, only allowing signals with a certain frequencyranges to pass the panel. This frequency range, i.e., those frequenciesthat are below the cutoff of C_(SIG) but above the cutoff of RC NetworksA and B, determines the stimulus frequencies that may be used to drivethe touch panel.

The panel will therefore impose a phase delay on the stimulus waveformpassing through it. This phase delay is negligible for traditionalopaque touch panels, wherein the electrode structure is typically formedby PCB traces, which have negligible resistance to their characteristicimpedance. However, for transparent panels, typically constructed usingIndium Tin Oxide (ITO) conductive traces, the resistive component may bequite large. This introduces a significant time (phase) delay in thepropagation of the stimulus voltage through the panel. This phase delaycauses the demodulation waveform to be delayed with respect to thesignal entering the pre-amplifier, thereby reducing the dynamic range ofthe signal coming out of the ADC.

To compensate for this phase delay, a delay clock register (“DCL”, notshown) may be provided, which can be used to delay the demodulationwaveform relative to the signal entering the preamplifier thereforecompensating for the external panel delay and maximizing the dynamicrange. This register is input into the demodulator 403 and simply delaysthe demodulation waveform by a predetermined amount. The amount may bedetermined either on startup of the panel by measurement, or may beestimated for the panel as a whole based on known manufacturingcharacteristics. Each pixel of the touch surface may have its ownuniquely determined delay parameter to fully optimize the readingcircuitry, or the delay parameter may be determined on a row by rowbasis. Adjustment would be generally similar to the techniques discussedabove for adjustment of the charge amplifier feedback capacitor and theoffset compensation voltage.

The demodulated signal is then passed to offset compensation circuitry.The offset compensation circuitry comprises mixer 402 and programmableoffset DAC 405. Mixer 402 takes the output voltage 453 of thedemodulator and subtracts an offset voltage (discussed below) toincrease the dynamic range of the system.

Offset compensation is necessary because the pixel capacitance C_(SIG)is comprised of a static part and a dynamic part. The static part is afunction of sensor construction. The dynamic part is a function of thechange of C_(SIG) when the finger approaches the pixel, and is thus thesignal of interest. The purpose of the offset compensator is toeliminate or minimize the static component thereby extending the dynamicrange of the system.

As noted above, the offset compensation circuitry is comprised of twoparts, a programmable offset DAC 405 and a mixer 402. Offset DAC 405generates a programmable offset voltage from the digital static offsetvalue VOFF_REG. This digital value is converted into a static analogvoltage (or current, if operating in the current domain) by the DAC andthen mixed (by mixer 403 b) with a voltage (or current) set by theabsolute value (determined by block 404 b) of the demodulation waveform.The result is a rectified version of the demodulation waveform, theamplitude of which is set by the static value of VOFF_REG and theabsolute portion of the demodulation waveform currently retrieved fromthe DMOD lookup table 404. This allows for the right amount of offsetcompensation for a given portion of the demodulation waveform. Thereforethe offset compensation waveform effectively tracks the demodulationwaveform.

As with the charge amplifier feedback capacitor, it is useful to adjustthe offset compensation circuitry to account for variations in theindividual pixel capacitance due to manufacturing tolerances, etc. Theadjustment may be substantially similar to that discussed above withrespect to the charge amplifier feedback capacitor. Specifically, theoffset voltage value stored in VOFF_REG may be calculated as follows:VOFF _(—) REG[Y]=VOFF _(—) UNIV+VOFF[Y]where Y is the individual column within a row, VOFF_UNIV is an offsetvoltage set on a row by row basis, and VOFF[Y] is a lookup table. Again,the adjustment could be performed on a true pixel by pixel basis orVOFF_UNIV could be a single constant value, depending on a particularimplementation. Also, although discussed in terms of rows and columns,the adjustment arrangement is equally applicable to non-Cartesiancoordinate systems.

As an alternative to the arrangement described above with respect toFIG. 10, the offset compensation could take place prior to demodulation.In this case, the shape of the offset compensation waveform has to matchthe waveform coming out of the preamplifier rather than the waveformcoming out of the demodulator, i.e., it has to be a square wave,assuming negligible attenuation in the panel, such that the shape of thedrive waveform is preserved. Also, if offset compensation is performedfirst, the offset waveform is an AC waveform with respect to thereference voltage, i.e., the maxima are positive in respect to V_(REF)and the minima are negative in respect to V_(REF). The amplitude of theoffset waveform is equivalent to the amount of offset compensation.Conversely, if demodulation is performed first, the offset waveform is aDC waveform, i.e. it is either positive in respect to Vref or negative(since the demodulated waveform is also DC in respect to Vref). Again,the amplitude in this case is equivalent to the amount of offsetcompensation for every part of the demodulated waveform. In essence, theoffset compensation circuit needs to correlate the amount of offsetcompensation needed dependent on the shape of the waveform.

The demodulated, offset compensated signal is then processed byprogrammable gain ADC 406. In one embodiment, ADC 406 may be asigma-delta, although similar type ADCs (such as a voltage to frequencyconverter with a subsequent counter stage) could be used. The ADCperforms two functions: (1) it converts the offset compensated waveformout of the mixer arrangement (offset and signal mixer) to a digitalvalue; and (2) it performs low pass filtering functions, i.e., itaverages the rectified signal coming out of the mixer arrangement. Theoffset compensated, demodulated signal looks like a rectified Gaussianshaped sine wave, whose amplitude is a function of C_(FB) and C_(SIG).The ADC result returned to the host computer is actually the average ofthat signal.

One advantage of using a sigma delta ADC is that such ADCs are much moreefficient for performing averaging in the digital domain. Additionally,digital gates are a lot smaller than analog low pass filters and sampleand hold elements, thus reducing the size of the total ASIC. One skilledin the art will further appreciated other advantages, particularly withregard to power consumption and clock speed.

Alternatively, one could use an ADC separate from the controller ASIC.This would require a multiplexor to share the ADC between multiplechannels and a sample and hold circuit for each channel to average andhold the average of the demodulation waveform. This would likely consumeso much die area as to be impractical for controllers intended for usewith touch surfaces having a large number of pixels. Additionally, toachieve acceptable operation, the external ADC would need to operatevery fast, as a large number of pixels must be processed very quickly toprovide timely and smooth results in response to a user's input.

As noted above, the sensor is driven at three different frequencies,resulting in three capacitance images, which are used for noiserejection as described below. The three frequencies are chosen such thatthe pass band at one particular frequency does not overlap with the passbands at the other frequencies. As noted above, a preferred embodimentuses frequencies of 140 kHz, 200 kHz, and 260 kHz. The demodulationwaveform is chosen such that the side bands are suppressed.

A standard sine wave, illustrated in FIG. 11A together with its passbandfrequency spectrum, may be used as a demodulation waveform. The sinewave provides a well-defined pass band with some stop band ripple.Alternatively, other waveforms having well defined pass bands withminimum stop band ripple could also be used. For example, aGaussian-enveloped sine wave, illustrated in FIG. 11B together with itspassband frequency spectrum, also has a well defined pass band, withless stop band ripple. One skilled in the art will appreciate that theshape and type of the demodulation waveform affects the passband of themixer, which, in turn, affects the effectiveness of the noisesuppression provided by the frequency hopping mechanism. As will befurther appreciated by those skilled in the art, other waveforms couldalso be used.

Turning now to FIG. 12, nine waveforms are illustrated that explain thenoise suppression features of the system. Voltage waveform 501 is asquare wave demonstrating the stimulus waveform applied to the sensor.Waveform 504 is the Gaussian enveloped sine wave signal used as ademodulation waveform. Waveform 507 is the output of the demodulator,i.e., the product of the waveforms 501 and 504. Note that it provides awell defined pulse at the fundamental frequency of the applied squarewave voltage.

The center column illustrates an exemplary noise waveform 502.Demodulation waveform 505 is the same as demodulation waveform 504. Notethat the demodulated noise signal 508 does not produce a significantspike, because the fundamental frequency of the noise signal is outsidethe passband of the demodulation signal.

The composite of the excitation waveform and noise signal is illustratedin 503. Again, demodulation waveform 506 is the same as demodulationwaveforms 505 and 504. The demodulated composite does still show thenoise waveform, although various signal processing algorithms may beapplied to extract this relatively isolated spike.

Additionally, noise rejection may accomplished by providing multiplestimulus voltage at different frequencies and applying a majority rulesalgorithm to the result. In a majority rules algorithm, for eachcapacitance node, the two frequency channels that provide the bestamplitude match are averaged and the remaining channel is disposed of.For example, in FIG. 13 vertical line 600 represents the measuredcapacitance, with the markings 601, 602, and 603 representing the threevalues measured at three stimulus frequencies. Values 602 and 603provide the best match, possibly suggesting that value 601 is corrupted.Thus value 601 is discarded and values 602 and 603 are averaged to formthe output.

Alternatively, a median filter could be applied, in which case value602, i.e., the median value would be selected as an output. As yetanother alternative, the three results could simply be averaged, inwhich case a value somewhere between value 601 and 602 would result. Avariety of other noise rejection techniques for multiple sample valueswill be obvious to those skilled in the art, any of which may suitablybe used with the controller described herein.

Operation of the circuit may be further understood with respect to FIG.14, which is a flowchart depicting operation of the controller. Oneskilled in the art will appreciate that various timing and memorystorage issues are omitted from this flowchart for the sake of clarity.

Image acquisition begins at block 701. The system then sets the clock soas to acquire samples at the middle clock frequency (e.g., 200 kHz) asdiscussed above with respect to FIG. 9 (block 702). The variousprogrammable registers, which control such parameters as voltage offset,amplifier gain, delay clocks, etc., are then updated (block 703). Allcolumns are read, with the result stored as a Mid Vector (block 704) Thehigh clock frequency is then set (block 705), and the steps of updatingregisters (block 706) and reading all columns and storing the result(step 707) are repeated for the high sample frequency. The clock is thenset to the low frequency (step 708) and the register update (block 709)and column reading (block 710) are repeated for the low samplefrequency.

The three vectors are then offset compensated, according to thealgorithm described above (block 711). The offset compensated vectorsare then subjected to a median filter as described above. Alternatively,the offset compensated vectors could be filtered by the majority rulesalgorithm described with respect to FIG. 13 or any other suitablefiltering technique. In any case, the result is stored. If more rowsremain, the process returns to the mid frequency sampling at block 702).If all rows are completed (block 713), the entire image is output to thehost device (block 714), and a subsequent new image is acquired (block701).

While this invention has been described in terms of several preferredembodiments, there are alterations, permutations, and equivalents, whichfall within the scope of this invention. For example, the term“computer” does not necessarily mean any particular kind of device,combination of hardware and/or software, nor should it be consideredrestricted to either a multi purpose or single purpose device.Additionally, although the embodiments herein have been described inrelation to touch screens, the teachings of the present invention areequally applicable to touch pads or any other touch surface type ofsensor. Furthermore, although the disclosure is primarily directed atcapacitive sensing, it should be noted that some or all of the featuresdescribed herein may be applied to other sensing methodologies. Itshould also be noted that there are many alternative ways ofimplementing the methods and apparatuses of the present invention. It istherefore intended that the following appended claims be interpreted asincluding all such alterations, permutations, and equivalents as fallwithin the true spirit and scope of the present invention.

1. A controller for a touch surface, the touch surface having aplurality of drive electrodes and at least one sense electrode, aplurality of nodes formed at intersections of the plurality of driveelectrodes and the at least one sense electrode, the controllercomprising: output circuitry operatively connected to the plurality ofdrive electrodes, the output circuitry being configured to generatetiming signals used to generate drive waveforms for the touch surface,each drive waveform including a plurality of bursts in a singlestimulation sequence for stimulating the drive electrodes in a singlescan of the nodes in the touch surface, the plurality of burstsincluding a first periodic waveform having a first predeterminedfrequency, and at least one additional periodic waveform having at leastone additional predetermined frequency different from the firstpredetermined frequency; and input circuitry operatively connected tothe at least one sense electrode, the input circuitry being configuredto determine proximity of an object at each node by measuring capacitivecoupling of the drive waveforms from the drive electrode to the senseelectrode of the node; wherein at least one of the drive electrodes isstimulated consecutively with the plurality of bursts including periodicwaveforms having different predetermined frequencies before one of theother drive electrodes is stimulated in the single scan.
 2. Thecontroller of claim 1 further comprising decoding and level shiftingcircuitry connected between the output circuitry and the driveelectrode, the decoding and level shifting circuitry being configured toreceive the timing signals and generate drive waveforms for the touchsurface.
 3. The controller of claim 2 wherein the decoding and levelshifting circuitry are part of the single application specificintegrated circuit.
 4. The controller of claim 1 wherein the at leastone additional periodic waveform comprises a second periodic waveformhaving a second predetermined frequency and a third periodic waveformhaving a third predetermined frequency, each of the second predeterminedfrequency and the third predetermined frequency being different from thefirst predetermined frequency and different from each other.
 5. Thecontroller of claim 1 wherein the input circuitry comprises a chargeamplifier, the charge amplifier further comprising: an operationalamplifier having an inverting input terminal, a non-inverting inputterminal, and an output terminal, wherein the inverting input terminalis operatively connected to the at least one sense electrode; a feedbackcapacitor connected between the output terminal and the inverting inputterminal, wherein the feedback capacitor is programmable to take on arange of values; and a feedback resistor connected between the outputterminal and the inverting input terminal, wherein the feedback resistoris programmable to take on a range of values.
 6. The controller of claim5 wherein the charge amplifier further comprises a resistor coupledbetween the inverting input terminal and the at least one senseelectrode to form an anti-aliasing filter in combination with thefeedback resistor and feedback capacitor.
 7. The controller of claim 5wherein the non-inverting input of the amplifier is coupled to ground.8. The controller of claim 1 wherein the input circuitry comprises anoffset compensator, the offset compensator comprising: a programmableoffset digital to analog converter adapted to generate an offset signalcorresponding to a static component of the capacitive coupling betweenthe drive electrode and the sense electrode; and a subtractor circuitconfigured to subtract the offset signal from a measured signalindicative of the capacitive coupling between the drive electrode andthe sense electrode.
 9. The controller of claim 5 wherein the inputcircuitry further comprises an offset compensator, the offsetcompensator comprising: a programmable offset digital to analogconverter adapted to generate an offset signal corresponding to a staticcomponent of the capacitive coupling between the drive electrode and thesense electrode; and a subtractor circuit configured to subtract theoffset signal from an output signal of the charge amplifier, the outputsignal being indicative of the capacitive coupling between the driveelectrode and the sense electrode.
 10. The controller of claim 1 whereinthe input circuitry comprises a demodulator, the demodulator comprisinga multiplier configured to mix a signal indicative of a capacitivecoupling between the drive electrode and the sense electrode with ademodulation waveform.
 11. The controller of claim 5 wherein the inputcircuitry further comprises a demodulator, the demodulator comprising amultiplier configured to mix an output signal of the operationalamplifier, said output signal being indicative of a capacitive couplingbetween the drive electrode and the sense electrode, with a demodulationwaveform.
 12. The controller of claim 11 wherein the input circuitryfurther comprises an offset compensator, the offset compensatorcomprising: a programmable offset digital to analog converter adapted togenerate an offset signal corresponding to a static component of thecapacitive coupling between the drive electrode and the sense electrode;and a subtractor circuit configured to subtract the offset signal fromthe output signal of the demodulator, said output signal beingindicative of the capacitive coupling between the drive electrode andthe sense electrode.
 13. The controller of claim 8 wherein the inputcircuitry further comprises a demodulator, the demodulator comprising amultiplier configured to mix an output signal of the offset compensator,said output signal being indicative of a capacitive coupling between thedrive electrode and the sense electrode, with a demodulation waveform.14. The controller of claim 10, wherein the demodulation waveform isdetermined with reference to a lookup table.
 15. The controller of claim14 wherein the demodulation waveform is a Gaussian-enveloped sine wave.16. The controller of claim 1, wherein the input circuitry furthercomprises an analog to digital converter configured to produce a digitaloutput from the measured capacitive coupling of the drive waveforms fromthe drive electrode to the sense electrode.
 17. The controller of claim16 wherein the analog to digital converter is a sigma-delta converter.18. A method of operating a touch surface, the touch surface comprisinga plurality of drive electrodes and at least one sense electrode, aplurality of nodes formed at intersections of the plurality of driveelectrodes and the at least one sense electrode, the method comprising:stimulating the drive electrodes with a plurality of bursts in a singlestimulation sequence in a single scan of the nodes in the touch surface,the plurality of bursts including a first periodic waveform having afirst predetermined frequency and at least one additional periodicwaveform having an additional predetermined frequency different from thefirst predetermined frequency; reading the at least one sense electrodeafter the drive electrodes have been stimulated with the first periodicwaveform during the single scan to determine a first capacitance of thenodes formed at the intersection of the drive electrodes and the atleast one sense electrode; reading the at least one sense electrodeafter the drive electrodes have been stimulated with an additionalperiodic waveform during the single scan to determine at least oneadditional capacitance of the node formed at the intersection of thedrive electrode and the at least one sense electrode; and combining thefirst capacitance with the at least one additional capacitance todetermine a capacitance of the node, wherein at least one of the driveelectrodes is stimulated consecutively with the plurality of burstsincluding periodic waveforms having different predetermined frequenciesbefore one of the other drive electrodes is stimulated in the singlescan.
 19. The method of claim 18 wherein stimulating the at least onedrive electrode with at least one additional periodic waveform having anadditional predetermined frequency different from the firstpredetermined frequency comprises: stimulating the at least one driveelectrode with a second periodic waveform having a second predeterminedfrequency; and stimulating the at least one drive electrode with a thirdperiodic waveform having a third predetermined frequency; wherein thesecond and third predetermined frequencies are different from the firstpredetermined frequency and different from each other.
 20. The method ofclaim 19 wherein combining the first capacitance with the at least oneadditional capacitance comprises taking an average of the capacitances.21. The method of claim 19 wherein combining the first capacitance withthe at least one additional capacitance comprises applying a majorityrules algorithm to the capacitances.
 22. The method of claim 19 whereincombining the first capacitance with the at least one additionalcapacitance comprises taking the median of the capacitances determinedby the first, second, and third stimuli.
 23. A method of operating atouch surface, the touch surface comprising a plurality of driveelectrodes and at least one sense electrode, a plurality of nodes formedat intersections of the plurality of drive electrodes and the at leastone sense electrode, the method comprising: stimulating the driveelectrodes with a plurality of bursts in a single stimulation sequencein a single scan of the nodes in the touch surface, the plurality ofbursts including a first drive waveform having a first predeterminedfrequency and at least one additional drive waveform having at least oneadditional predetermined frequency different from the firstpredetermined frequency; detecting a first waveform on the at least onesense electrode caused by capacitive coupling of the first drivewaveform at the nodes; detecting at least one additional waveform on theat least one sense electrode caused by capacitive coupling of the atleast one additional drive waveform at the nodes; amplifying thedetected waveforms; demodulating each of the first waveform and the atleast one additional waveform; and determining a capacitance at thenodes to detect an object located proximate the nodes, wherein at leastone of the drive electrodes is stimulated consecutively by the pluralityof bursts including periodic waveforms having different predeterminedfrequencies before one of the other drive electrodes is stimulated inthe single scan.
 24. The method of claim 23, further comprisingsubtracting an offset from the amplified waveforms, the offset beingdetermined as a function of the capacitance of the nodes, wherein thedemodulating of the amplified waveform takes place subsequent tosubtracting the offset.
 25. The method of claim 23, wherein demodulatingeach of the first waveform and the at least one additional waveformcomprises mixing the waveforms with a Gaussian enveloped sine wave. 26.The method of claim 23, wherein demodulating each of the first waveformand the at least one additional waveform further comprises delaying ademodulating signal by a predetermined amount to compensate for phasedelay in the touch panel.
 27. The method of claim 23, further comprisingadjusting programmable elements of detecting and amplifying circuitry bypredetermined amounts to compensate for node-to-node variations.
 28. Themethod of claim 18, wherein combining includes performing one ofapplying a majority rules algorithm, selecting a median value, oraveraging the capacitive coupling.